As discussed in the background section of the above-referenced '163 patent, a wide variety of electronic circuit applications employ one or more transconductance stages to generate output/drive currents that can be reasonably accurately controlled for delivery to one or more loads. As a non-limiting example, various equipments employed by telecommunication service providers contain what are known as ‘SLIC’s (subscriber line interface circuits), to interface (transmit and receive) telecommunication signals with respect to (tip and ring leads of) a (copper) wireline pair.
Because the length of the wireline pair can be expected to vary from installation to installation, and may have a very significant length (e.g., on the order of multiple miles), and the wireline pair transports both substantial DC voltages, as well as AC signals (e.g., voice and/or ringing), designing a SLIC that has ‘universal’ use in both legacy and state of the art installations continues to be a daunting task for the circuit designer.
In order to accommodate the above-referenced parameter variations in a telecommunication signalling environment, it is customary practice to configure the SLIC as a transconductance amplifier-based circuit, that produces a prescribed output current in response to an input voltage. One of the issues involved in using a transconductance amplifier circuit is the fact that it must not only deliver a very precisely controlled output current, but must do so irrespective of the voltages of the supply rails from which it is powered.
Prior to the invention disclosed in the '163 patent, conventional transconductance amplifier stages, whether they involved single ended architectures or differentially coupled transistor pairs (such as that those shown at Q1–Q2 in FIG. 1), usually suffered from the presence of one or more non-linearities associated with unequal or mismatched diode junctions in the components of the circuit generating a single ended output current.
One way to obviate this problem has been to employ a differentially balanced operational amplifier circuit architecture, such as that diagrammatically illustrated in FIG. 2. As shown therein, a pair of operational amplifiers A1 and A2 may be coupled to respective drive inputs (bases) of a pair of transistors Q1–Q2. Transistors Q1 and Q2 have their output (collector-emitter) current flow paths coupled in a differential configuration between a current mirror circuit M and negative feedback paths of the amplifiers A1 and A2, which terminate opposite ends of an impedance (resistance) Z. Although such a dual amplifier circuit design enables an output current to be precisely generated in terms of an applied input voltage, it does so at an increase in complexity and therefore device count, power and cost, and is constrained by the large signal bandwidth limitations of the operational amplifiers.
Auspiciously, the bipolar transistor-based transconductance amplifier disclosed in the above-referenced '163 patent effectively remedies shortcomings of such conventional transconductance amplifier circuit designs, by transforming a single ended input voltage (which may be a composite of plural input voltages) into a very precise, single ended output current, yet without requiring a substantial quiescent current, and in a manner that is effectively independent of (differential) voltage supply rails through which the circuit is powered.
The architecture of the bipolar transistor-based transconductance amplifier circuit of the '163 patent is schematically shown in FIG. 3 as including an operational amplifier 100 configured as a unity gain buffer. Amplifier 100 has a dual polarity input operational amplifier input and gain stage 110, and a low output impedance, single ended output stage 120. The input stage 110, which may have a conventional high impedance, moderate voltage gain circuit configuration, has a first, non-inverting polarity input 111, that is coupled to a DC reference voltage, shown as a voltage v1 (relative to ground (GND)), and a second, inverting polarity input 112, which is coupled to the output 123 of the amplifier's output stage 120 by way of a negative feedback path 126. The reference voltage v1 can be selected in compliance with the overhead voltages and power dissipation required by the specific application in which the transconductance amplifier circuit is employed.
The output stage 120 includes a differentially coupled bipolar transistor circuit pair, having a first, diode-connected NPN transistor 130, whose collector 131 and base 132 are connected in common to a first polarity output port 113 of the amplifier's input stage 110. The emitter 133 of transistor 130 is coupled in common to the emitter 143 of a second, diode-connected PNP transistor 140. In a complementary fashion, PNP transistor 140 has its collector 141 and base 142 connected in common to a second polarity output port 114 of the amplifier input stage 110. The base 132 of NPN transistor 130 is coupled in common with the base 152 of an NPN transistor 150, the emitter 153 of which is coupled in common to the emitter 163 of a PNP transistor 160 and to an input/output node 123 of output stage 120.
The PNP transistor 160 has its base 162 coupled in common with the base 142 of the PNP transistor 140. The output stage's input/output node 123 is coupled over negative feedback path 126 to the inverting input 112 of the input stage 110. As noted above, unlike a conventional amplifier circuit, the input/output node 123, rather than being employed to supply an output current to a downstream load, is coupled to receive one or more input currents, respectively supplied through one or more coupling resistors Z1, . . . ZN, to associated voltage feed ports 125-1, . . . , 125-N.
The series-connected, collector-emitter current paths through the output transistors 150 and 160 of output stage 120, rather than being biased via a direct coupling to respective (Vcc and Vee) voltage supply rails 155 and 165, are coupled in circuit with first current supply paths 171 and 181 of first and second bipolar transistor-implemented current mirror circuits 170 and 180, respectively. These current mirror circuits serve to isolate the biasing of the amplifier's output stage 120 from its power supply terminals, so that the output current produced at a single ended output port 135 can be accurately controlled independent of the values of the power supply voltages.
The current mirror circuit 170 includes a first PNP transistor 200 having its emitter 203 coupled to the (Vcc) voltage supply rail 155, and its base 202 coupled in common with the base 212 and collector 211 of a diode-connected current mirror PNP transistor 210, whose emitter 213 is coupled to (Vcc) voltage supply. rail 155. The current mirror transistor 200 supplies a mirrored output current to the current supply path 172 as a prescribed factor K of the current received by transistor 210 over the current supply path 171, in accordance with the ratio (1:K) of the geometries of the transistors 210/200. The collector 211 and base 212 of transistor 210 are coupled over the first current supply path 171 of the current mirror 170 to the collector 151 of transistor 150 of output stage 120. The collector 201 of transistor 200 is coupled over a second current supply path 172 of the current mirror 170 to a transconductance stage output node 135.
In a complementary manner, current mirror circuit 180 includes a first NPN transistor 220 having its emitter 223 coupled to the (Vee) voltage supply rail 156 and its base 222 coupled in common with the base 232 and collector 231 of a diode-connected current mirror NPN transistor 230, whose emitter 233 is coupled to (Vee) voltage supply rail 156. The collector 231 and base 232 of the current mirror transistor 230 are coupled over the first current supply path 181 of current mirror 180 to collector 161 of output stage transistor 160. The collector 221 of transistor 220 is coupled over a second current supply path 182 of the current mirror 180 to the output node 135. The current mirror transistor 220 provides a mirrored output current to current supply path 182 as a factor K of the current received by transistor 230 over current supply path 181, in accordance with the (1:K) ratio of the geometries of transistors 230/220.
An examination of current node equations that define the transfer function of the bipolar process-based transconductance amplifier circuit of FIG. 3, reveals that it has a very wide dynamic range that not only accommodates multiple, differential polarity voltages applied at its voltage feed ports 125-1, . . . , 125-N, but enjoys very low quiescent power dissipation.
More particularly, the single ended output current i123 delivered to input/output node 123 may be defined in equation (1) as:
                                                                        i                123                            =                                                                    (                                                                  v                                                  125                          -                          1                                                                    -                                              v                        111                                                              )                                    /                                      Z                    1                                                  +                                  …                  ⁢                                                                          ⁢                                                            (                                                                        v                                                      125                            -                            N                                                                          -                                                  v                          111                                                                    )                                        /                                          Z                      N                                                                                                                                              =                                                ∑                                      i                    =                    1                                    N                                ⁢                                                      (                                                                  v                                                  125                          -                          i                                                                    -                                              v                        111                                                              )                                    /                                      Z                    i                                                                                                          (        1        )            
The currents i171 and i181 supplied to current mirrors 170 and 180 may be related to the current i123 at the input/output node 123 by equation (2) as:i123+i171=i181==>i123=181−i171  (2)
The currents i172 and i182 supplied by current mirrors 170 and 180 may be related to the current i135 at the output node 135 by equation (3):i172+i135=i182  (3)
and equation (4) as:Ki171+i135=Ki181=>iout=135=K(i181−i171)=Ki123  (4)
Substituting equation (1) into equation (4) yields equation (5) as:
                              i          out                =                  K          ⁢                                    ∑                              i                =                1                            N                        ⁢                                          (                                                      v                                          125                      -                      i                                                        -                                      v                    111                                                  )                            /                              Z                i                                                                        (        5        )            
Implicit in equations (2) and (4) is the fact that transistor limitations due to beta and early voltage are compensated or minimized (in a manner not specifically shown in the diagrammatic illustration of FIG. 3). It may also be noted that if transistors 130/150 and 140/160 are matched pairs and the time average value of each of the input voltages applied to the voltage input terminals 125-1, . . . , 125-N is equal to v1, and v1 is a DC voltage, then the time average values of currents i171 and i181 are equal to the DC bias current IDC flowing in the emitter path of the output stage transistors 130-140. Therefore, if the value of the bias current IDC is relatively low and the current mirror ratio K is equal to or less than 1, the quiescent power consumed by the transconductance amplifier circuit of FIG. 3 can be reduced to a very small value.
As further described in the '163 patent, a particularly useful application of the transconductance amplifier circuit of FIG. 3 is as a building block for one or more subcircuits employed within a subscriber line interface circuit, or SLIC, for interfacing communication signals supplied from a device such as a modem, with a wireline pair for delivery to a remote circuit, such as a subscriber's telephone. To this end, the front end of the SLIC's receiver channel circuit has a transconductance circuit that incorporates the amplifier of FIG. 3 as its basic building block. It also includes a pair of auxiliary current mirror circuits, which are cross-coupled with the current mirror circuits 170 and 180 of the transconductance amplifier of FIG. 3.
The current node relationships associated with this cross-coupling of these auxiliary current mirror circuits with current mirror circuits 170 and 180 are such that the auxiliary current mirror circuits supply to an additional output port the same precision output current, but in an opposite current flow directional sense, that is provided at the output port of the transconductance amplifier of FIG. 3. These two opposite polarity current output ports are applied through respective ‘tip’ and ‘ring’ output amplifiers, which are coupled in a voltage follower configuration to respective tip and ring output ports for application to a (telephone) wireline pair being driven by the receiver channel circuit. For additional details of the incorporation and operation of the transconductance amplifier of FIG. 30 in SLIC applications, attention may be directed to the '163 patent.